1. Field
Embodiments generally relate to the field of current feedback instrumentation amplifiers.
2. Background
Instrumentation amplifiers are commonly used to amplify small differential input voltages while rejecting common-mode input voltages. A desired feature of such amplifiers is a low input-referred offset voltage combined with a low input current. The latter can be achieved by using a MOS input stage, but such an input stage typically results in a high offset voltage.
Another desired feature of instrumentation amplifiers is that their input range includes the negative supply rail, so that they can be connected to a grounded signal source in a single-supply system. This is not possible with a conventional 3-opamp instrumentation amplifier topology.
This limitation to some extent has been overcome by using a current-feedback topology with PMOS input transistors. The PMOS transistors transfer the differential input voltage to a resistor connected between their sources, resulting in a current proportional to the differential input voltage. The PMOS transistors at the same time provide the required common-mode level-shift to be able to make this voltage-to-current conversion with an input voltage at ground level. In the rest of the amplifier, the generated current is converted back into an output voltage using a second resistor.
FIG. 1 shows a block diagram of one conventional current-feedback instrumentation amplifier 100. The differential input voltage Vin is converted to a current by transconductance amplifier g2. As described above, amplifier g2 has PMOS input transistors which enable it to sense input signals at the negative supply rail. The difference between the output voltage Vout and a reference voltage Vref is scaled down by a resistive divider, consisting of R1 and R2, to provide a feedback voltage Vfb. This is applied to a second transconductance amplifier g3. The feedback loop, closed by the output stage g1, ensures that the output current of g3 equals that of g2. It is appreciated that the output stage, here shown as a single Miller-compensated transconductance stage, can in practice consist of multiple stages. If the two transconductances g2 and g3 are equal, Vfb equals Vin, and therefore the output voltage equals:Vout=Vref+(R1+R2)/R2·Vin  (1)
In the more general case that the two transconductances are not equal, the output voltage equals:Vout=Vref+g2/g3·(R1+R2)/R2·Vin  (2)
In addition to its ability to sense input voltages at the negative supply rail, amplifier 100 has the attractive feature that its output can swing rail-to-rail, which is important in low-voltage applications.
However, circuit 100 is disadvantageous in that the offsets of transconductance amplifiers g2 and g3 add directly to the input voltage, and therefore need to be compensated for. FIG. 2 illustrates one conventional amplifier 200 that employs chopper switches 210 and 220 added at the input of g2 and g3 to periodically reverse the polarity of the input and feedback signal. An additional chopper switch 230 at the input of the output stage restores the original polarity. This configuration effectively modulates the offset of the transconductance amplifiers to the chopper frequency, where it can, in principle, be filtered out.
An important disadvantage of using chopping to eliminate the offset in current-feedback instrumentation amplifiers is that the modulated offset results in spurious AC signals at the output of the amplifier 200. For example, the output of amplifier 200 may actually appear as a sawtooth signal. Since the output of an instrumentation amplifier is typically sampled by an analog-to-digital converter, such spurious signal may result in measurement errors unless they are filtered out. Conventional implementations have attempted to reduce and filter these spurious signals by using a continuous (non-chopped) feedforward path and various extra offset-compensation loops. This, however, leads to a very large and complex system.
Another important disadvantage of using chopping is that the input source is exposed to a switched capacitive load consisting of the input capacitance Cin2 of transconductance amplifier g2. Due to the periodic polarity reversal, this input capacitance has to be alternately charged to +Vin and −Vin. The associated current results in an input offset current. Effectively, this reduces the input impedance of the instrumentation amplifier (e.g., amplifier 200) to:Rin=1/(2·fchop·Cin2).  (3)For typical values of fchop=10 kHz and Cin2=1 pF, the input impedance is 50 MΩ. In contrast, non-chopped instrumentations amplifiers with MOS inputs typically achieve input impedances on the order of 10 GΩ. This reduced impedance due to chopping can cause significant gain errors when reading out a high-impedance signal source. A similar problem occurs at the input of transconductance amplifier g3, whose input capacitance Cin3 presents a switched load to the feedback network.
Thus, conventional current feedback instrumentation amplifiers do not provide a simple way to reduce input offsets while at the same time maintaining high input impedance and avoiding spurious signals at the output.